AD600ARZ-R7 datasheet(18/28 Pages) AD

18 / 28 page

AD600/AD602

Rev. E | Page 18 of 28

A simple half-wave detector is used based on Q1 and R2. The

average current into Capacitor C2 is the difference between the

current provided by the AD590 (300 μA at 300 K, 27°C) and the

collector current of Q1. In turn, the control voltage VG is the

time integral of this error current. When VG (thus the gain) is

stable, the rectified current in Q1 must, on average, balance

exactly the current in the AD590. If the output of A2 is too

small to do this, VG ramps up, causing the gain to increase until

Q1 conducts sufficiently. The operation of this control system

follows.

First, consider the particular case where R2 is zero and the

output voltage VOUT is a square wave at, for example, 100 kHz,

well above the corner frequency of the control loop. During the

time VOUT is negative, Q1 conducts. When VOUT is positive, it is

cut off. Since the average collector current is forced to be

300 μA and the square wave has a 50% duty-cycle, the current

when conducting must be 600 μA. With R2 omitted, the peak

value of VOUT would be just the VBE of Q1 at 600 μA (typically

about 700 mV) or 2 VBE p-p. This voltage, thus the amplitude at

which the output stabilizes, has a strong negative temperature

coefficient (TC), typically –1.7 mV/°C. While this may not be

troublesome in some applications, the correct value of R2

renders the output stable with temperature.

To understand this, first note that the current in the AD590 is

closely proportional to absolute temperature (PTAT). In fact,

this IC is intended for use as a thermometer. For the moment,

assume that the signal is a square wave. When Q1 is conducting,

VOUT is the sum of VBE. VOUT is also a voltage that is PTAT and

that can be chosen to have a TC equal but opposite to the TC of

the base-to-emitter voltage. This is actually nothing more than

the band gap voltage reference principle in thinly veiled

disguise. When R2 is chosen so that the sum of the voltage

across it and the VBE of Q1 is close to the band gap voltage of

about 1.2 V, VOUT is stable over a wide range of temperatures,

provided Q1 and the AD590 share the same thermal environment.

Since the average emitter current is 600 μA during each half-

cycle of the square wave, a resistor of 833 Ω would add a PTAT

voltage of 500 mV at 300 K, increasing by 1.66 mV/°C. In

practice, the optimum value of R2 depends on the transistor

used and, to a lesser extent, on the waveform for which the

temperature stability is to be optimized; for the devices shown

and sine wave signals, the recommended value is 806 Ω. This

resistor also serves to lower the peak current in Q1 and the

200 Hz LP filter it forms with C2 helps to minimize distortion

due to ripple in VG. Note that the output amplitude under sine

wave conditions is higher than for a square wave because the

average value of the current for an ideal rectifier would be

0.637 times as large, causing the output amplitude to be 1.88 V

(= 1.2/0.637), or 1.33 V rms. In practice, the somewhat nonideal

rectifier results in the sine wave output being regulated to about

1.275 V rms.

An offset of 375 mV is applied to the inverting gain-control

inputs C1LO and C2LO. Therefore, the nominal –625 mV to

+625 mV range for VG is translated upwards (at VG´) to –0.25 V

for minimum gain to +1 V for maximum gain. This prevents

Q1 from going into heavy saturation at low gains and leaves

sufficient headroom of 4 V for the AD590 to operate correctly

at high gains when using a 5 V supply.

In fact, the 6 dB interstage attenuator means that the overall

gain of this AGC system actually runs from –6 dB to +74 dB.

Thus, an input of 2 V rms would be required to produce a

1 V rms output at the minimum gain, which exceeds the 1 V rms

maximum input specification of the AD600. The available gain

range is therefore 0 dB to 74 dB (or X1 to X5000). Since the gain

scaling is 15.625 mV/dB (because of the cascaded stages), the

minimum value of VG´ is actually increased by 6 × +15.625 mV,

or about 94 mV, to −156 mV, so the risk of saturation in Q1 is

reduced.

The emitter circuit of Q1 is somewhat inductive (due its finite ft

and base resistance). Consequently, the effective value of R2

increases with frequency. This results in an increase in the

stabilized output amplitude at high frequencies, but for the

addition of C3, determined experimentally to be 15 pF for the

2N3904 for maximum response flatness. Alternatively, a faster

transistor can be used here to reduce HF peaking. Figure 38

shows the ac response at the stabilized output level of about

1.3 rms. Figure 39 demonstrates the output stabilization for the

sine wave inputs of 1 mV to 1 V rms at frequencies of 100 kHz,

1 MHz, and 10 MHz.

FREQUENCY (MHz)

100

0.1

3dB

Figure 38. AC Response at the Stabilized Output Level of 1.3 V rms

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