AD829

Data Sheet

Rev. I | Page 12 of 20

THEORY OF OPERATION

The AD829 is fabricated on the Analog Devices, Inc., proprietary

complementary bipolar (CB) process, which provides PNP and

NPN transistors with similar fTs of 600 MHz. As shown in

Figure 35, the AD829 input stage consists of an NPN differential

pair in which each transistor operates at a 600 µA collector current.

This gives the input devices a high transconductance, which in

turn gives the AD829 a low noise figure of 2 nV/√Hz at 1 kHz.

Figure 35. Simplified Schematic

The input stage drives a folded cascode that consists of a fast pair of

PNP transistors. These PNPs drive a current mirror that provides a

differential-input-to-single-ended-output conversion. The high

speed PNPs are also used in the current-amplifying output stage,

which provides a high current gain of 40,000. Even under heavy

loading conditions, the high fTs of the NPN and PNPs, produced

using the CB process, permit cascading two stages of emitter

followers while maintaining 60 phase margin at closed-loop

bandwidths greater than 50 MHz.

Two stages of complementary emitter followers also effectively

buffer the high impedance compensation node (at the CCOMP pin)

from the output so that the AD829 can maintain a high dc open-

loop gain, even into low load impedances (92 dB into a 150 Ω

load and 100 dB into a 1 kΩ load). Laser trimming and PTAT

biasing ensure low offset voltage and low offset voltage drift,

enabling the user to eliminate ac coupling in many applications.

For added flexibility, the AD829 provides access to the internal

frequency compensation node. This allows users to customize the

frequency response characteristics for a particular application.

Unity-gain stability requires a compensation capacitance of 68 pF

(Pin 5 to ground), which yields a small signal bandwidth of

66 MHz and slew rate of 16 V/µs. The slew rate and gain

bandwidth product varies inversely with compensation

capacitance. Table 4 and Figure 37 show the optimum

compensation capacitance and the resulting slew rate for

a desired noise gain.

For gains between 1 and 20, choose CCOMP to keep the small signal

bandwidth relatively constant. The minimum gain that will still

provide stability depends on the value of the external

compensation capacitance.

An RC network in the output stage (see Figure 35) completely

removes the effect of capacitive loading when the amplifier

compensates for closed-loop gains of 10 or higher. At low

frequencies, and with low capacitive loads, the gain from the

compensation node to the output is very close to unity. In this case,

C is bootstrapped and does not contribute to the compensation

capacitance of the device. As the capacitive load increases, a pole

forms with the output impedance of the output stage, which

reduces the gain, and subsequently, C is incompletely

bootstrapped. Therefore, some fraction of C contributes to the

compensation capacitance, and the unity-gain bandwidth falls.

As the load capacitance is further increased, the bandwidth

continues to fall, and the amplifier remains stable.

EXTERNALLY COMPENSATING THE AD829

The AD829 is stable with no external compensation for noise

gains greater than 20. For lower gains, two different methods of

frequency compensating the amplifier can be used to achieve

closed-loop stability: shunt and current feedback compensation.

SHUNT COMPENSATION

Figure 36 and Figure 37 show that shunt compensation has an

external compensation capacitor, CCOMP, connected between the

compensation pin and ground. This external capacitor is tied in

parallel with approximately 3 pF of internal capacitance at the

compensation node. In addition, a small capacitance, CLEAD, in

parallel with resistor R2, compensates for the capacitance at the

inverting input of the amplifier.

Figure 36. Inverting Amplifier Connection Using External Shunt

Compensation

Figure 37. Noninverting Amplifier Connection Using External Shunt

Compensation

Table 4 gives the recommended CCOMP and CLEAD values, as well

as the corresponding slew rates and bandwidth. The capacitor

values were selected to provide a small signal frequency response

with <1 dB of peaking and <10% overshoot. For Table 4, ±15 V

–IN

+IN

1.2mA

OFFSET NULL

C

12.5pF

R

500

Ω

OUTPUT

+VS

–VS

CCOMP

15

Ω

15

Ω

2

3

7

6

4

–

+

AD829

+VS

–VS

0.1

µF

0.1

µF

5

CCOMP

50

Ω

50

Ω

COAX

CABLE

R1

CLEAD

R2

VOUT

1k

Ω

VIN

3

2

7

6

4

+

–

AD829

+VS

–VS

0.1

µF

0.1

µF

CLEAD

5

CCOMP

50

Ω

50

Ω

CABLE

VOUT

R2

R1

1k

Ω

VIN