SC4503 datasheet(13/22 Pages) SEMTECH

13 / 22 page

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SC4503

POWER MANAGEMENT

Applications Information (Cont.)

circuit from the driving logic gate during fault condition.

In Figure 5(f) the shutdown pin is driven from a logic gate

whose V

OH is higher than the supply voltage to the SC4503.

The diode clamps the maximum shutdown pin voltage to

one diode voltage above the input power supply.

During soft-start, C

SS is charged by the difference between

the R

SS current and the shutdown pin current,

. In

steady state, the voltage drop across R

SS reduces the shut-

down pin voltage according to the following equation:

−

(14)

In order for the SC4503 to achieve its rated switch current,

must be greater than 2V in steady state. This

puts an upper limit on R

SS for a given enable voltage VEN (=

voltage applied to R

SS). The maximum speciﬁed

µA with

(see “Electrical Characteristics”).

The largest R

SS can be found using (14):

−

If the enable signal is less than 2V, then the interfacing

options shown in Figures 5(d) and 5(e) will be preferred. The

methods shown in Figures 5(a) and 5(c) can still be used

however the switch current limit will be reduced. Variations

and switch current limit with

SHDN SS

SHDN

pin voltage

and temperature are shown in the “Typical Characteristics”.

Shutdown pin current decreases as temperature increases.

Switch current limit at a given

also decreases as

temperature rises. Lower shutdown pin current ﬂowing

through R

SS at high temperature results in higher shutdown

pin voltage. However reduction in switch current limit (at

a given

) at high temperature is the dominant

effect.

Feed-Forward Compensation

Figure 6 shows the equivalent circuit of a boost converter.

Important poles and zeros of the overall loop response

are:

Low frequency integrator pole,

−

Output ﬁlter pole,

−

Compensating zero,

−

and

Right half plane (RHP) zero,

()

−

()

−

The poles p

1, p2 and the RHP zero z2 all increase phase

shift in the loop response. For stable operation, the over-

all loop gain should cross 0dB with -20dB/decade slope.

Due to the presence of the RHP zero, the 0dB crossover

frequency should not be more than

. The internal

compensating zero z

1 provides phase boost beyond p2. In

general the converter is more stable with widely spaced

ﬁlter pole p

2 and the RHP zero z2. The RHP zero moves to

low frequency when either the duty-cycle D or the output

current I

OUT increases. It is beneﬁcial to use small inductors

and larger output capacitors especially when operating at

high

ratios.

A feed-forward capacitor C

4 is needed for stability. The value

of C

4 can be determined empirically by observing the induc-

tor current and the output voltage during load transient.

Starting with a value between

µµ

and

µµ

, C

4 is

adjusted until there is no excessive ringing or overshoot in

inductor current and output voltage during load transient.

Sizing the inductor such that its ripple current is about 0.5A

also improves phase margin and transient response.

POWER

STAGE

REFERENCE

VOLTAGE

1.252V

COMP

ESR

VOUT

VIN

I OUT

RO is the equivalent output resistance of the error amplifier

POWER

STAGE

REFERENCE

VOLTAGE

1.252V

COMP

ESR

VOUT

VIN

I OUT

RO is the equivalent output resistance of the error amplifier

Simpliﬁed Equivalent Model of a Boost

Converter

Figure 6. Simpliﬁed Equivalent Model of a Boost

Converter

Figure 6.

POWER

STAGE

REFERENCE

VOLTAGE

1.252V

COMP

ESR

VOUT

VIN

I OUT

RO is the equivalent output resistance of the error amplifier

POWER

STAGE

REFERENCE

VOLTAGE

1.252V

COMP

ESR

VOUT

VIN

I OUT

RO is the equivalent output resistance of the error amplifier

Simpliﬁed Equivalent Model of a Boost

Converter

Figure 6. Simpliﬁed Equivalent Model of a Boost

Converter

Figure 6.

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