LT1336CS datasheet(11/16 Pages) LINER

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LT1336

APPLICATIONS INFORMATION

Using the components as shown in Figure 2 the flyback

regulator will run at around 800kHz. To lower the fre-

quency CFILTER can be increased and to increase the

frequency CFILTER can be decreased.

Power MOSFET Selection

Since the LT1336 inherently protects the top and bottom

MOSFETs from simultaneous conduction, there are no

size or matching constraints. Therefore, selection can be

made based on the operating voltage and RDS(ON) require-

ments. The MOSFET BVDSS should be at least equal to the

LT1336 absolute maximum operating voltage. For a maxi-

mum operating HV supply of 60V, the MOSFET BVDSS

should be from 60V to 100V.

The MOSFET RDS(ON) is specified at TJ = 25°C and is

generally chosen based on the operating efficiency re-

quired as long as the maximum MOSFET junction tem-

perature is not exceeded. The dissipation in each MOSFET

is given by:

PD I

DS ON

() +

() ( )

∂

where D is the duty cycle and

∂ is the increase in RDS(ON)

at the anticipated MOSFET junction temperature. From

this equation the required RDS(ON) can be derived:

DS ON

( ) = () +()

∂

For example, if the MOSFET loss is to be limited to 2W

when operating at 5A and a 90% duty cycle, the required

RDS(ON) would be 0.089Ω/(1 + ∂). (1 + ∂) is given for each

MOSFET in the form of a normalized RDS(ON) vs tempera-

ture curve, but

∂ = 0.007/°C can be used as an approxima-

tion for low voltage MOSFETs. Thus, if TA = 85°C and the

available heat sinking has a thermal resistance of 20

°C/W,

the MOSFET junction temperature will be 125

°C and

∂ = 0.007(125 – 25) = 0.7. This means that the required

RDS(ON) of the MOSFET will be 0.089Ω/1.7 = 0.0523Ω,

which can be satisfied by an IRFZ34 manufactured by

International Rectifier.

Transition losses result from the power dissipated in each

MOSFET during the time it is transitioning from off to on,

or from on to off. These losses are proportional to (f)(HV)2

and vary from insignificant to being a limiting factor on

operating frequency in some high voltage applications.

Figure 2. Using the Flyback Regulator

The flyback regulator works as follows: when switch S is

on, the primary current ramps up as the magnetic field

builds up. The magnetic field in the core induces a voltage

on the secondary winding equal to V +. However, no power

is transferred to VBOOST because the rectifier diode D2 is

reverse biased. The energy is stored in the transformer’s

magnetic field. When the primary inductor peak current is

reached, the switch is turned off. Energy is no longer

transferred to the transformer causing the magnetic field

to collapse. The collapsing magnetic field induces a change

in voltage across the transformer’s windings. During this

transition the Switch pin’s voltage flies to 10.6V plus a

diode above V +, the secondary forward biases the rectifier

diode D2 and the transformer’s energy is transferred to

VBOOST. Meanwhile the primary inductor current goes to

zero and the voltage at ISENSE decays to the lower inductor

current threshold with a time constant of (RSENSE)(CFILTER),

thus completing the cycle.

SWITCH

SV+

PV+

RSENSE

Ω

1/4W

1N4148

40V

1N4148

24V

1000pF

6.2k

HV =

60V MAX

1336 F02

LT1336

T1*

1:1

1N4148

* COILTRONICS CTX100-1P

SWGND

ISENSE

CBOOST

µF

CFILTER

0.1

µF

–

VBOOST

TGATEDR

TGATEFB

BOOST

TSOURCE

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