ADN8834ACPZ-R2 datasheet(20/27 Pages) AD

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ADN8834

Data Sheet

Rev. B | Page 20 of 27

MOSFET DRIVER AMPLIFIERS

The ADN8834 has two separate MOSFET drivers: a switched

output or pulse-width modulated (PWM) amplifier, and a high

gain linear amplifier. Each amplifier has a pair of outputs that drive

the gates of the internal MOSFETs, which, in turn, drive the TEC as

shown in Figure 33. A voltage across the TEC is monitored via

the SFB and LDR pins. Although both MOSFET drivers achieve

the same result, to provide constant voltage and high current,

their operation is different. The exact equations for the two

outputs are

VLDR = VB − 40(VOUT2 − 1.25 V)

VSFB = VLDR + 5(VOUT2 − 1.25 V)

where:

VOUT2 is the voltage at OUT2.

VB is determined by VVDD as

VB = 1.5 V for VVDD < 4.0 V

VB = 2.5 V for VVDD > 4.0 V

The compensation network that receives the temperature set voltage

and the thermistor voltage fed by the input amplifier determines

the voltage at OUT2. VLDR and VSFB have a low limit of 0 V and

an upper limit of VVDD. Figure 37, Figure 38, and Figure 39 show

the graphs of these equations.

OUT2 (V)

1.25

0.75

0.25

1.75

2.25

2.75

–2.5

2.5

7.5

5.0

VSYS = 5.0V

VSYS = 3.3V

Figure 37. LDR Voltage vs. OUT2 Voltage

OUT2 (V)

1.25

0.75

0.25

1.75

2.25

2.75

–2.5

2.5

7.5

5.0

VSYS = 5.0V

VSYS = 3.3V

Figure 38. SFB Voltage vs. OUT2 Voltage

–2.5

–5.0

2.5

5.0

OUT2 (V)

VSYS = 5.0V

VSYS = 3.3V

1.25

0.75

0.25

1.75

2.25

2.75

Figure 39. TEC Voltage vs. OUT2 Voltage

PWM OUTPUT FILTER REQUIREMENTS

A type three compensator internally compensates the PWM

amplifier. As the poles and zeros of the compensator are designed

and fixed by assuming the resonance frequency of the output

LC tank being 50 kHz, the selection of the inductor and the

capacitor must follow this guideline to ensure system stability.

Inductor Selection

The inductor selection determines the inductor current ripple and

loop dynamic response. Larger inductance results in smaller

current ripple and slower transient response as smaller inductance

results in the opposite performance. To optimize the performance,

the trade-off must be made between transient response speed,

efficiency, and component size. Calculate the inductor value

with the following equation:

(

)

OUT

∆

–

where:

VSW_OUT is the PWM amplifier output.

fSW is the switching frequency (2 MHz by default).

∆IL is the inductor current ripple.

A 1 µH inductor is typically recommended to allow reasonable

output capacitor selection while maintaining a low inductor current

ripple. If lower inductance is required, a minimum inductor value

of 0.68 µH is suggested to ensure that the current ripple is set to

a value between 30% and 40% of the maximum load current,

which is 1.5 A.

Except for the inductor value, the equivalent dc resistance (DCR)

inherent in the metal conductor is also a critical factor for

inductor selection. The DCR accounts for most of the power loss

on the inductor by DCR × IOUT2. Using an inductor with high

DCR degrades the overall efficiency significantly. In addition,

there is a conduct voltage drop across the inductor because of

the DCR. When the PWM amplifier is sinking current in cooling

mode, this voltage drives the minimum voltage of the amplifier

higher than 0.06 × VIN by at least tenth of millivolts. Similarly, the

maximum PWM amplifier output voltage is lower than 0.93 × VIN.

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