AD9853_15 datasheet(17/31 Pages) AD

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OBSOLETE

AD9853

–17–

REV. C

the next contains the message information and is used to de-

modulate the signal instead of the absolute phase of the signal.

The transmitter and receiver must use the same symbol deriva-

tion scheme.

Differential encoding in the AD9853 occurs while data still

exists as a serial data stream. When in straight QPSK or 16-QAM,

the serial data stream passes to the symbol mapper/format en-

coder stage without modification. When differential encoding is

engaged, the serial data stream is modified prior to the symbol

mapper/format stage according to Table VI. Only I1 and Q1 are

modified, even in the D16-QAM mode whose symbols are com-

posed of Q1, I1, Q0, I0. In D16-QAM, only the two MSBs of

the 4-bit symbol are modified; furthermore, the “previously

transmitted symbol” referred to in Table VI are the two MSBs

of the previous 4-bit symbol.

Symbol mapping for QPSK and DQPSK are identical. Symbol

mapping for 16-QAM and D16-QAM are slightly different (see

Figure 37) in accordance with MCNS (DOCSIS) specifications.

Special Note: For most modulation modes, a minimum pre-

amble is required. For DQPSK the minimum preamble is one

symbol (2 bits) and for either 16-QAM or D16-QAM the mini-

mum preamble is one symbol (4 bits). For FSK or QPSK, no

preamble is required.

User should be additionally aware that in the DQPSK mode,

the preamble is not differentially encoded in accordance with

MCNS (DOCSIS) specifications. If the preamble must be dif-

ferentially encoded, it can “pre-encoded” using the derivation in

Table VI. In D16-QAM, the preamble is always differentially

encoded as is the “payload” data.

When initiating a new differentially encoded transmission, the

“previously transmitted symbol” is always the last symbol of the

preamble.

PROGRAMMABLE PULSE-SHAPING FIR FILTERS

The I and Q data paths of the modulator each contain a pulse

shaping filter. Each is a 41-tap, linear phase FIR. They are used

to provide bandwidth containment and pulse shaping of the data

in order to minimize intersymbol interference. The filter coeffi-

cients are programmable, so any realizable linear phase response

characteristic may be implemented. The linear phase restriction

is due to the fact that the user may only define the center coeffi-

cient and the lower 20 coefficients. The hardware fills in the

upper 20 coefficients as a mirror image of the lower 20. This

forces a linear phase response. It should also be noted that the

pulse shaping filter upsamples the symbol rate by a factor of

four.

Normally, a square-root raised cosine (SRRC) response is desired.

In fact, the AD9853 Evaluation Board software driver implements

an SRRC response. When using the SRRC response, an excess

bandwidth factor (

α) is defined that affects the low pass roll-off

characteristic of the filter (where 0

≤ α ≤ 1). When α = 0, the

SRRC is an ideal low-pass filter with a “brick wall” at one-half

of the symbol rate (the Nyquist bandwidth of the data). Although

this provides maximum bandwidth containment, it has the ad-

verse affect of causing the tails of the time domain response to

be large, which increases intersymbol interference (ISI). On the

other hand, when

α = 1, the SRRC yields a smooth roll-off

characteristic that significantly reduces the time domain tails,

which improves ISI. Unfortunately, the cost of this benefit is a

doubling of the bandwidth of the data signal. Values of

α between

0 and 1 yield a tradeoff between excess bandwidth in the fre-

quency domain and tail suppression in the time domain.

The FIR filter coefficients for the SRRC response may be calcu-

lated using a variety of methods. One such method uses the

Inverse Fourier Transform Integral to calculate the impulse re-

sponse (time domain) from the SRRC frequency response (fre-

quency domain). An example of this method is shown in Figure

33. Of course, this method requires that the SRRC frequency

response be known beforehand.

The FIR filters in the AD9853 are implemented in hardware

using a fixed point architecture of 10-bit, twos complement

integers. Thus, each of the filter coefficients, ai, is an integer

such that:

–512

≤ a

≤ 511

[i = 0, 1, … , 40]

PROGRAMMABLE INTERPOLATION FILTERS

The AD9853 employs two stages of interpolation filters in each

of the I and Q channels of the modulator. These filters are

implemented as Cascaded Integrator-Comb (CIC) filters. CIC

filters are unique in that they not only provide a low-pass fre-

quency response characteristic, but also provide the ability to

have one sampling rate at the input and another sampling rate at

the output. In general, a CIC filter may either be used as an

interpolator (low-to-high sample rate conversion) or as a

decimator (high-to-low sample rate conversion). In the case of

the AD9853, the CIC filters are configured as interpolators,

only. Furthermore, the interpolation is done in two separate

stages with each stage designed so that the rate change is pro-

grammable. The first interpolator stage offers rate change ratios

of 3 to 31, while the second stage offers rate change ratios of 2

to 63.

As stated in the previous section, the data coming out of the

FIR filters is oversampled by four. Spectral images appear at

their output (a direct result of the sampling process). These

images are replicas of the baseband spectrum which are re-

peated at intervals of four times the symbol rate (the rate at

which the FIR filters sample the data). The images are an un-

wanted byproduct of the sampling process and effectively repre-

sent a source of noise.

Normally, the output of the FIR filters would be fed directly to

the input of the I and Q modulator. This means that the spectral

images produced by the FIRs would become part of the modu-

lated signal—definitely not a desirable consequence. This is

where the CIC filters play their role. Since they have a low-pass

characteristic, they can be used to eliminate the spectral images

produced by the FIRs.

Frequency Response of the CIC Filters

The frequency response of a CIC filter is predictable. It can be

shown that the system function of a CIC filter is:

()













−

∑

Where N is the number of cascaded integrator (or comb) sec-

tions, R is the rate change ratio, and M is the number of unit

delays in each integrator/comb stage. For the AD9853, two of

these variables are fixed as a result of the hardware implementa-

tion; specifically, N = 4 and M = 1. As mentioned earlier, R (the

rate change ratio) is programmable.

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